Radar signal simulator



Jan. 19, 1960 c. H. McsHAN RADAR SIGNAL SIMULATOR 7 sheets-sheet 1 Filed March 30. 1954 qu l C. H. MCSHAN RADAR SIGNAL SIMULATOR Filed Mar'ch so, 1954 F/E. E.

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l RADAR SIGNAL SIMULATOR FiledMaroh 30. 1954 7 Sheets-Sheet 3 f-f/fi. 4.

INVENTOR. CLARENCE Hun-fera ma SHAH Jan. 19, 1960 c. H. McsHAN RADAR SIGNAL sIMuLAToR 7 Sheets-Sheet 4 Filed March 30, 1954 l I l NWN INVENTOR CLARE/yc; HUNTER mc SHA/'7 v- ,v A, f A

' Jan. 19, 1960 c. H. MGSHAN Y 2,922,157

RADAR SIGNAL SIMULATOR FiledV March 30, 1954 7 Sheets-Sheet 5 FAC-T7. 7.

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,Pz/55o JWM 3 l INVENTOR. CLARENCE HUNTER Pk SHAn fw/lfw Jan. 19, 1960 Filed` March 30 1954 C. H. MCSHAN RADAR SIGNAL SIMULATOR '7 Sheets-Sheet G INVENTOR. CLARE/ice Ha/1ER Mc SHAP/ Jan. 19, 1'960 c. H. McsHAN RADAR SIGNAL SIMULATOR '7 Sheets-Sheet 7 Filed MarCh 30, 1954 2 2 m W a, M lo INVENTOR. Cl-H REP/CE Hur/75K ma SHAP/ y BY afg(

RADAR SIGNAL SllVIULATOR Clarence H. McShan, Great Neck, N.Y., assgnor to Pitometer Log Corporation, New York, N.Y., a corporation of New York Application March 30, 1954, Serial No. 419,660 s claims. (ci. 34a-17.7)

The present invention relates to a test set for simulating radar echo signals and more particularly to a test set for simulating doppler-modulated echo signals.

It is a general object of the invention to provide a test set which is capable of measuring various operational characteristics of a radar unit of the moving target type (MTI).

A more specific object of the invention is to provide a test set capable of measuring sub-clutter visibility, that is, the strength of a moving target signal, relative to the fixed target clutter, which can just be detected and recognized as a moving target on a radar indicator.

Another more specific object of the invention is to provide a test set which may be used to check the accuracy of the radar range markers.

Still another more specific object of the invention is to provide a test set which may be used to view the shape of the transmitted radar signal and the shape of the simulated radar signals.

Another specific object of the invention is to provide apparatus for producing simulated doppler-modulated echo signals, the degree of doppler-modulation of which can be varied.

Another specific object of the invention is to provide a means of checking the output carrier frequency of a radar unit.

Another object of the invention is to provide an improved balanced modulator capable of attenuating received radio frequency pulses, detecting said received radio frequency pulses, and heterodyning intermediatefrequency pulses with the output of a local oscillator to produce simulated radio-frequency echo pulses.

A further object of the invention is to provide a novel phase modulation apparatus.

Still another object of the invention is to provide a new and improved delay line arrangement.

With the above objects in view, the present invention mainly consists of a radar signal simulator including means adapted to be coupled to the radar unit it is desired to test, for receiving therefrom carrier frequency radar pulses. Means are provided in the simulator for producing at least one retarded carrier frequency pulse in synchronism with each of the received radar pulses, respectively, and for shifting the phase of adjacent retarded carrier frequency pulses by a predetermined amount while maintaining alternate ones of the retarded carrier frequency pulses in phase.

In a preferred embodiment of the invention a plurality of carrier frequency pulses are produced in synchronism with each received carrier frequency radar pulse. These pulses are retarded given fixed amounts relative to one another. Adjacent groups of said retarded pulses are shifted in phase a given amount while alternate groups of said retarded carrier frequency pulses are maintained in phase.

In the preferred embodiment of the invention the received radio frequency radar pulses are converted to intermediate frequency pulses prior to the shifting 2,922,157 Patented Jan. 19, 1,960

' 2 in phase thereof. The phase shifting is produced by a novel phase modulating device comprising a cathode follower means having a capacitive load impedance across which the output signal is taken.y The intermediate frequency retarded pulses are supplied to the cathode follower means and means are provided for changing the transconductance of the cathode follower in accordancewith a given parameter of a modulating signal. This 'causes the output signal of the cathode follower means to be phase modulated in accordance with the foregoing parameter of the modulating signal.

The novel features which are considered as characteristic for the invention are set forth in particular in the appended claims. The invention itself, however, both as to its construction and its method of operation, together with additional objects and advantages thereof, will be best understood from the following ldescription of specific embodiments when read in connection with the accompanying drawings, in which:

Fig. 1 is a block diagram of a signal simulator in accordance with the invention;

Fig. 2 is a diagram, partically in cross-section and partially schematic, of the balanced modulator of the signal simulator; l

Fig. 3 is a schematic diagram of the radio-frequency equivalent circuit of the balanced modulator illustrated in Fig. 2; t v

Fig. 4 is a schematic diagram of an intermediate-frequency equivalent circuit of the balanced modulator illustrated in Fig. 2;

Fig. 5 is a schematic diagram of a portion of the signal simulator illustrated in Fig. 1 including the doppler modulator, phase shift arrangement and various other circuit components;

Fig. 6 is a schematic diagram of the first intermediatefrequency stage of the signal simulator;

Fig. 7 shows the waveforms present at various points in the signal simulator;

Fig. 8 is a drawing of a plan-position oscilloscope display of xed targets simplated by the' signal simulator;

Fig. 9 is a drawing of a plan-position oscilloscope display of pulsed moving targets. simulated by -thesignal simulator; Y Y

Fig. l0 isa diagram of the phase shifter showngin block form in Fig. 1; and A y Fig. ll is a vector diagram relating to the dopplermodulator shown in Fig. 5. f

In the drawing like reference numerals refer to like elements.

Referring now to the drawing and more particularly to Fig. l, there is shown a block diagram of the radar signal simulator including a balancedv mixer 20 which is directly coupled to the directional coupler of a radar unit under test. (It is to be understood that the inventionis equally applicable to the case where the balanced mixer.

is not directly coupled to the radar unit but is instead provided With antenna means or a pick-up loop of some other type for receiving a transmitted radar pulse.) Balanced mixer 2t) attenuates the received radio-frequency pulses to a usable level. The output of the balanced mixer 20 and the output of the stable local-oscillator 21 are heterodyned in crystal mixer 22 to produce intermediate-fre-v 3 modulator 24 which shifts the phase of adjacent intermediate-frequency radar pulses as will be explained more fully below. This adds a moving target component to the simulated vecho p ulses. 'Iheoutput of doppler-modulator24is supplied to intermediate-frequency'amplifiers 25,26, and the output of second intermediate-frequencyl amplifier 26 isl fed to a delay means 27 and thence back to the input of the first amplifier 25. The intermediatefrequency radar pulses circulate through theA loop comprising blocks 25, 26 and 27 accumulating on each cycle a given time delay. v constructed, a delay of 61.8 microseconds wasintroduced by the delay means 27 so that the simulated echo pulses were spaced apart the equivalentwof 5' nautical miles. After a predetermined number of circulations. through loopd2'5, 2,6, 27, agating pulse lfrom one-shot multivibrator 31r is fed to the control gridV ofthefirst IgEamplier 25' through crystal rectifier, 3 5 which terminatesn the circnlationvofpulses through the loop'.A Inrthe signal simulatot huilt, or more simulated intermediate-frequency pulses were produced in response to eaclrreceivedy radar Pulsev The output of'third intermediate-frequency amplifier 28 is applied through adjustable attenuator 291to Vthe balanced/mixer where th'e trains of intermediate-fre-u quency 'pulses are mixed with the output of stable localoscillator 21 and converted to trains of simulated radiofrequency target pulses. The latter are applied directly to the directional coupler' of the radar unit undertestY through the same lead as the radar input pulses arereceived. Of course, if an antenna is used, instead of being coupled directly to the radar unit, thesimulated target pulses will be radiated toward the radary unit and the latter will'receive these pulses on its ownrantenna.

The output of third intermediate-frequency amplilier 28 is also supplied to monitor detector 36 which .is coupledtocathode follower 37. At point 38, there appears a video output ofthe simulated intermediate-frequency target pulses and these pulses may be viewed by connecting a test oscilloscope to 38. Y

Elements 30-33 comprise the signal simulator timing circuits. These perform a number of'functions including gating of `intermediate-frequency amplier 2S' to provide a desired plurality of simulated intermediate echo pulses in synchronism with each received'radar radio frequency pulse; biasing of doppler-modulator 24 to produce phase shifts in the simulated moving targets; and pulsing action in order to facilitate measurement of sub-clutter visibility. o Y r In brief, the timing circuits operate as follows: The balanced mixer 20 supplies detected radar pulses to pulse amplifier 30 which in turn triggers one shot multivibrator 31. Inrits quiescent condition, one half of multivibrator 31 conducts and the other half is cut-ntf. During its quiescent state, a steady bias is applied to the crystal rectifier 35 so that it conducts and eiectively shunts the control grid4 to ground and thereby prevents the amplifier fromV amplifyingY any further signals applied by delay means 27'. Upon the application of a pulse tomultivibrator 31, the latter generates a negative-going gate which is applied to rectifier 35 causing said rectifier to be cutfoff and-thereby to providea high impedance `between the controlgrid of amplifier and ground. During the negative gating interval, signals supplied tothe control gridv by delaymeans 27 are amplified. In a preferred embodiment of the in vention, the negative gating interval is made s uiciently long to permit 10 or more simulated echo pulses to be produced in synchronism. withV each received radar pulse. The above gatingaction will'be describedin somewhat more detail ybelow in connection with Fig. 6;

Flip-Hop triggerk 32Uis a circuit having .two conditions o f stable equilibrium.

the invention this trigger circuit comprises a multivibrator one-half of which conducts during one ot," the In the signalrsimulator actllally.V

In ,a preferredY embodiment of` conditions of stable equilibrium and the other half of whichfconducts during the other of the conditions of stable equilibrium. The circuit remains in either Yof the two conditions, with no change in plate, grid or cathode potential, until some action occurs which causes the nonconducting section to conduct. The tube sections then reverse their functions and remain in the new condition until plate current flows in the cut-off section.

Flips-floptrigger 32 is triggered by one-shot multivibrator 31. Upon being triggered, circuit 32 generates a negative going square Wave pulse inV its output circuit which is applied to the control grid of doppler modulator 24 when doppler switch 34 is in certain positions.

The time relationship-of the square wave outputs of one-shot multivibrator 31 and flip-flop trigger 32 is shown in Fig. 7, waveform 181 being the output of one-shot multivibrator 31 and waveform 185 the output of flipflop trigger. 32. As can be seen,rthe flipt-Hop trigger is actuatedV upon each occurrence of a positive-going leading edge184 of Waveform 181. The frequency output of the Hip-flop.trigger circuit' is one-half that of the oneshot multivibrator.

Free running multivibrator 33 is placedin circuit in` one position of the doppler switch 34. Inthis position, the cathode load impedance of the free running multi- `vibrator 32 is'` reduced and the; multivibrator, oscillates at a predetermined rate gating the flip-flop trigger 32 at this'rate.V In an lembodiment of the invention actually constructed, the free running multivibrator 33 had a symmetrical output wave at a frequency of aboutS cycles per second.

ReferringV now to Fig. 2,v there is shown av diagram, partially in` y cross section and partially schematic, of the balanced mixer 20. It comprises aY Conductive shell 49 Within which are mounted diodes 511 and 51. The diodes include tubular metal portions 75; 76 and 75a, 76a, respectively, and glass sections 77 and 77a, respectively. Radio-frequency radar input pulses aresupplied through coaxial line 69 to sleeves 70, 71 and thence to the respective cathodes 52, 53. These sleeves act as distributed inductance elements las will be illustrated below. As can be seen, the input signals are applied in push-pull. The tube outputsV at anodes 54 and 55, respectively, are fed through secondary windings 62 andy 63respectively, of 'intermediate frequency pulse transformer 74 and iiltercoil 67' and capacitor 6,8', rto coaxial line72.

Attennated radio-frequency pulses are coupled' by means ofloop 79 to coaxial line 516, and ,thence tothe applied from coaxial. line-73, to. theprimary winding 64 of vintermediate-frequency transformer 74 4and thence through windings 62, 63 to plates 54 and 55,; Primary winding 64 is grounded gto shell 49 through RC lter lnetwork 6,566' AThe intermediate-frequency pulses are heterodyned in the. balancedmixer toy produce simulated radio-frequency target pulses at the radarrfrequency. The simulated target pulses are taken from` cathodes S2, 53 and fed 'through coaxial line 69 back to the directional coupler of the radar unit under test. Y

The radio-frequency.operation of balanced mixer 20 will be more readily understood by referring to Fig. 3 which is the' radio-frequency equivalent circuit of the mixer. The balanced mixerris essentially a slightly unbalanced bridgeA circuit. Radiol frequency inputV signals from the radar-unit` under test are applied to the cathodes 52, 5,3 of the diodes 50, 51 in push-pull through distributed inductances 82, 83. These distributed inductances are provided by the circular` sleeves 70, 7,1 (Fig. 2) in which the diodes are mounted. Inl the test set actually built, a pair of type 6173 pulse diodes were employed for the balanced modulator tubes, however, it is to be uudrgstood that other types of tubes or nonlinear devices such as crystals are usable for the same purpose. The outputs of the balanced mixer. tubes are supplied through capacitors 84 and 85 respectively to the cavity sections 86, and 87 respectively represented in the schematic diagram as tuned circuits. These cavity sections are provided by the hollow metal sections of the diodes as shown in Fig. 2. The attenuated output is taken between ground 81 and the junction 80 of the two resonant cavity sections. As can be seen from Figs. 1 and 2, the attenuated radio-frequency output pulses are supplied through coaxial line 56 to mixer 22 where the pulses are heterodyned with the output of stable local oscillator 21 to produce intermediate frequency pulses.

If the bridge circuit were perfectly balanced, the radiofrequency potential existing at the junction 80 of the two cavity sections would be exactly equal to that appearing at the junction 81 of the tube cathode inductances 82, 83. Under this condition, the radio-frequency output would be zero. Since, in practice, a slight unbalance always does exist, a radio-frequency potential difference also exists between points 80 and 81. In the test set built, it was found that the power level of the output radio-frequency pulses was about 25 db below the level of the input radio-frequency pulses (ratio of 316 to 1). As already mentioned, the attenuated radio-frequency output of the balanced mixer is carried by coaxial cable 56 to a junction point with the output of stable local oscillator 21. As shown in Fig. 1, this junction point is one-quarter wavelength removed from the stable local oscillator output circuit and this causes a very high impedance to be seen by the radar radio-frequency pulses looking into oscillator 21vand oscillator 21 is thus isolated from the output of the balanced mixer.

Fig. 4 is the intermediate-frequency equivalent circuit of the balanced mixer. Intermediate-frequency pulses from the intermediate frequency circuits are supplied to the primary winding 64 of intermediate-frequency transformer 74, through secondary windings 62, 63, respectively, of said intermediate frequency transformer and thence to the respective plates 54, 55 of diodes 50, 51. The output of the stable local oscillator 21 is supplied through coaxial line 56 to the respective anodes 54 and 55 of the balanced mixer. As can be seen, the balanced mixers are fed in parallel by the stable local oscillator. Capacitors 90, 91 represent capacitors 61, 58, 57 and 60, 59, 57, respectively (see Fig. 2). The intermediate-frequency pulses and the output of the stable local oscillator are heterodyned in balanced mixer 50, 51 producing simulated radio-frequency target pulses at precisely the frequency of the original radar radio-frel quency input. These simulated radio-frequency target pulses are applied to the radar directional coupler through coaxial cable 69 (see Fig. 2).

The stable local oscillator circuit is not shown in detail since numerous arrangements known in the art may be employed. In the embodiment of the invention constructed, a lighthouse tube was used as the oscillator tube in a grounded-grid, coaxial line type arrangement. The output frequency of the stable local oscillator was made adjustable in the range of about 1190 to 1340 megacycles. This frequency band was required for the so-called L radar band which is 1215-1355 megacycles. It should be understood that the inventive concept is equally applicable to other frequencies and that in the case of higher radar frequencies other types of oscillator arrangements such as, for example, klystron oscillators or magnetron oscillators may be advantageously employed.

In the test set built, mixer stage 22 was a crystal mixer, however, other types of mixers known in the art may be employed instead. I

Referring now to Fig. 5, there is shown a schematic diagram of phase shifter 23, doppler modulator 24, doppler switch 34, and portions of timing stages 31, 32 and 33.

Phase shifter 23 is a variable artificial transmission line which introduces a phase shift in the intermediate-- frequency pulse output of mixer 22 prior to the application of the intermediate frequency pulses to dopplermodulator 24. The phase shifter is shown in greater detail in Fig. 10 which should also be referred to in the discussion which follows. As can be seen in Fig. 10, fixed coil 106 comprises a helix which is bent to have a circular `axis and which is wound of many turns of tine wire. Axial strips of copper 221 provide distributed shunt capacitance which, combined with the distributed inductance of the coil, make the unit into a transmission line whose characteristic impedance in the embodimen-t actually 'built was about 500 ohms. The pick-up coil 105 is wound coaxially with the delay line and is adjustable in position so that the time of arrival of intermediate frequency pulses at the pick-up coil depends upon the distance of the pickup coil from the input end (the end fed by mixer 22) of fixed coil 106. By varying the position of the pick-up coil, the signal delay can be varied up to about 0.029

microsecond, which is equivalent to'an 180 phase shift at 17.5 megacycles, the intermediate frequency employed. The movable coil may be adjusted in position by turning a hand crank 107 which is mechanically coupled to the moving coil through a mechanical coupling shown schematically by the dashed line 220. The phase shifter may be calibrated by providing the hand crank with a pointer which is movable relative to a fixed scale 222. In the model of the test set constructed, the phase shifter was calibrated at 5 intervals.

It is to be understood that more conventional types of phase Shifters may be employed instead of the one described above. These conventional arrangements include a fixed artificial delay line and a conductive wiper which is movable along the length of the delay line. An important advantage of the arrangement described above is the almost total absence of noise as the movable coil is moved relative to the xed coil. Another important advantage is that there is no danger that poor conductive connections will ever develop. Another advantage of the described arrangement is its infinite phase resolution. A nal advantage of the described arrangement is that there is no touching of parts and therefore phase adjustments may be made smoothly and accurately.

Referring again to Fig. 5, it is seen that the intermediate frequency signal output of the phase shifter 23 is fed through coupling capacitor and resistor 126 to the control grid 109 of the doppler-modulator tube 108. The doppler modulator varies the phase angle of the intermediate frequency signal. In the Off-position 119 of doppler switch 34, the bias signal supplied to control grid 109 of tube 108 remains the same and therefore the phase angle of successive intermediate frequency pulses also remains the same. In the Steady position 120 of switch 34, a square wave signal (Fig. 7) is applied to the control grid 109 which causes a phase shift between adjacent trains of intermediate frequency pulses. The amount of phase shift depends upon the amplitude of the square wave signal which in turn depends upon the values of resistors 112f-119 and the setting of potentiometer arm 138. In the Pulsed position 121 of doppler switch 34, free running multivibrator 33 is actuated and the action of doppler modulator 24 alternates between the Off and Steady conditions at a rate of about 5 cycles per second. In the 100% position 122 of doppler switch 34, the Output of iiip-llop trigger 32 is applied withl less attenuation to the control grid of the doppler modulator 24 and cuts said modulator off and therefore the train of intermediate frequency pulses on alternate pulse periods.

Doppler modulator 24 is a cathode-follower with a capacitive load 128. As already mentioned, intermediate-frequency pulses from phase shifter 23 are applied to the control grid 109 of the doppler modulator. In operation, the phase shifter 23 is adjusted to provide maximum phase detector response of the MTI radar receiver. This control, of course, is effective only for simulated moving target-s; Adjustable, capacitor 127 tunes'.

' are effected by varying. the resistive component, that is,

the tube resistance.` This tube resistancel is Varied by changing its grid bias .periodically by means of the output square wave offilip-ilop trigger 32'. Y

The quiescent' operating condition of doppler modulator 24 and alsoits condition;` of iminim'urrr` internal tube impedance are set by-fa combination. ofr'cathode and grid bias. The cathode-bias resistoroff the dopplermodulator 24 is resistor 130 and the latters- D-.C. return path is through coil 124 and movable coil V105'to ground. In the test set constructed theV minimum internaltube impedance was adjusted to provide a 27 phase angle shift relative to the phase of the applied intermediate Vfrequency pulses. In adjusting thevdoppler modulator to its quiescent condition, switch arm 133 is thrown to contact 135'which position will subsequently be termed the calibrate '1 position. In this position a small positive voltagevisftappedoif the cathode bias resistor 131 of the third 'intermediatefrequency amplier28 and fed through resistors 140,V 137 and 126 to the control grid 109 of thedopplermodulator. The value of the voltage tapped-olf determinesl the phase angle imparted to the intermediate frequency pulses -during the quiescent condition of the doppler modulator.-

After thelabove adjustment is made, switch arm 133 is thrown back to position 134 which will hereafter be termed the calibrate 2 position. t f

In operation, assume-that doppler switch 34 is in. the Steady position 120. A square-wave from the plate V159 of flip-flop trigger 32'is generated in the vplate-load resistance network consisting of resistors 112119.y The square wave is taken from arm 138 offpotentiornet'er 117 and appliedl through arm 123,' coupling capacitor 139 and resistors 137 and 126 to control grid 10911 The direct current component of the square wave isV removed by condenser. 139. The most positive portion of the resulting square wave'is clamped by diode 136 atV the quiescent bias voltageset by resistor 131. The negative portion of V-the square wave swings the grid of the' doppler modulator negativecausing an increase in its internal tube impedanceand a change. in the output signal phase angle. The amplitude of theV square wave, which determines the extent of phase angle change, may be varied by varying the position of arm 138. In theY X position, the amplitude of the square wave is maximum and in the Y position it is minimum. Accordingly, when'arm 138 is in the X position, maximum phase angle change results and when in theV Y position minimum `phase angle change results. Since the greater the change. in phase angle the greater the intensity of thev simulated doppler target signals as viewed on the cathode rayV oscilloscope of the MTI: radar unit under test, it is convenient to calibrate potentiometer 1-17 in terms of decibels. In the test set constructed the values of the resistor elements in the outputnework'112-119 wereso chosen 4that there was obtained at the X position a phasel shift corresponding to -10 .db and' at the Y'position a'phase shift corresponding. to 40 db. This Vtest set alsoincluded a scale calibrated in decibels.:so. thatphaset-shifts'Y corresponding to values vbetween -l'db and '-40.db could readily beobtained.

forms 188 and 189. Since the bias applied to control grid 109 by the flip-flop trigger 32 is diierent during adjacent puls'e'periods (see waveform 185, Fig. 7)', the amplitudes of succeedingpulse trains, 188, 189'are dierent. ,Since'the MTIA radar under test includesV means for limiting'allreceived'pulses to Vthe same amplitude, this is' of no consequence. There is a shift in phase ofthe intermediate carrier frequency of pulses 188 :with respect to the phase of the intermediate carrier frequency of pulses 189, thev extent of shift depending on the grid bias applied to grid 109. In the unit built, inthe -10 db position of switch arm 138, the phase angle changes on succeeding pulse periods from 27 to 62 and back again for a phase shift of 35 andA in the -40 db position of switch arm 138; the phase shift was approximately 1.

In the Pulsedjposition of doppler switch- 34 the second bank of the switch is thrown to positionv 121a. Inthis position the normal relatively high cathodeload resistor 157 of free running multivibrator 33 is shunted by resistor 158 which, isV of relativelyv low value. In. the embodiment of the test set: built resistor'157 was 100,000 ohms and resistor 158, 1,800 ohms. As shown in Fig. l, the output of free running multivibrator 33. is coupled to flip-flop trigger 32 L and causes theV flip-Hop trigger to be cut oit 5 times per second. The output waveform-192 of the free running multivibrator is shown in Fig. 7'. It includes a positive-going half cycle' 193 which isAO of a-second long and a-negative-going. half cycle-194 which is also a tenth of a secondv long.` During' the positivegoing half cycle 193, the output of the .doppler-1 modulator-` isV thesamey as when the doppler switch is in-V the-'steadyy position as indicated by successive wave--trains 1952 (Fig. 7). During the negative-going halfV cycle,:-thel.free running multivibratori causes theip-op-triigger tosupply a sufficiently negative signal to theV doppler modulator to cut the latter 01T. Accordingly, during-this `intervalgcorresponding to interval 196 (Fig. 7), there are noI intermediate-frequency pulses produced and: tl'l'erefo're-Y no simulated.radio-frequencyy target pulses produced. The use of the Pulsed simulated signals will be discussedbelow. l i In the position 122 of doppler-switch.' 34,?the square wave outputrof the flip-flop trigger is taken'frornthe junction of resistors 118 and 119.L Accordingly, negative-going portions of the square wave'are of-much greater amplitude than in the Steady or Pulsed'positions of switch 34. The resistor values are' so chosen-that the square wave is sufficiently negative to cut-oifdopper modulator 24 during alternate pulse periods.l The-resultant output 191, y19161 ofthe doppler modulator is shown in Fig. 7. Since in the radar unit under testl the phase of pulses in succcssivetrains is compared, it canreadily. be seen that there is maximum doppler-response in the 100% position. Y -v p vAs already mentioned, in the Oif position-of* doppler switch 34 there is no phase shift between successive trains of simulatedtarget pulses. The resultant trains ofoutput pulses are shown in Fig. 7 as waveforms 190-'190c.

These pulses are simulative of fixed targets:

`The theory of operation of the doppler modulator can more easily be understood by referring to the vectorV (l) 0=arc tan Xe The R-AR factoris 'set by theresistiveoutputrimpede ance of doppler modulator 23, which. depends vupon 'theV transconductance, R-AR=1/gm. (This is approximate;-

the exact value is n+1/rp.) The Xc value is that of condenser 128 plus the shunting stray capacitances to ground. The values of Xn and l/gm are chosen so as to set to equal 27, as shown in Fig. 1l. In the test set constructed, a gm value of 4500 micromhos was chosen and condenser 128 has a value of 21.2 micromicrofarads (including stray capacitances, making the output resistance of tube Y108,218.4 ohms. However, for the sake of mathematical simplicity, in the discussion which follows a gm value of 4000 micromhos and a capacitance value of 18 micro-microfarads are used. The theoretical discussion is, of course, applicable to any eX- ample.

When the gm value is set at 4000 micromhos:

frequency of 17.5 mc. gives:

(2) Xc=500 ohms ?Ll. o (3) tan 0-500-2, 9-27 The doppler modulator tube has its quiescent transconductance set at 4,000 micromhos during the fcalibrate l setting, as will be explained below. This value represents both the quiescent value, and the operating maximum value, since in operation, the tube grid is pulsed negative from its quiescent bias. This negative grid drive brings the transconductance of the doppler modulator to, a new lower value, and the tube cathode output resistance which is 250 ohms, represented in Fig. 11 by R-AR now rises to the higher value R, which correspondsito the new and lower transconductance of the doppler modulator during the period of the negative grid pulse. The vector a, which was the non-doppler modulated signal, with components Xc and R-AR, now shifts to the new position shown as vector b, whose phase angle corresponds to the component Xc arid R. The change of the resistant component R-AR to R shifts the signal phase through the angle 9, which, as shown in Fig. 11, is the angle between vector a and vector b. If the signals are limited to the same amplitude in the radar receiver or if means, not shown in any of the figures, are provided in the test set itself for limiting the amplitude of the signals, then both doppler and nondoppler modulated vectors can be taken as being equal, as shown in Fig. 11. It can be shown that the magnitude with which the vectors are assumed limited does not affect the sub-clutter visibility as long as complete limiting does occur and both doppler and non-doppler signals are limited to equal amplitudes. In the case of limited signals, the tip of the received vector will describe a circle C, shown in Fig. l1 to have a radius of length b, which is here taken as the limiting amplitude. When the doppler modulation is such that the vector output from the generator shifts through the angle A0, then the MTI radar receives a signal whose center, or average position, is along vector A, the angle bisector of A0. Vector A is a fictitious vector to the mid-point of the line joining the tips of vectors b and ka, where k is the scalar amplification required to bring the length of vector a equal to vector b.

The MTI radar acts as though it were receiving a signal whose xed component is vector A, with a doppler component vector S which has constant amplitude, but changes in phase to give the vectors b and ka. An extended discussion of this action may be found in the volume Radar System Engineering, by Ridenour, chapter 16. Assuming that there is maximum response velocity so that vector ka represents the target return from one transmitted radar pulse, vector b is the return from the next pulse, and the residue, or difference between successive return pulses, has a magnitude 2S. This is equivalent to assuming that the residue vector S shifts phase by 180 during successive pulse periods. The

sub-clutter visibility can be defined as the ratio of half the residue amplitude to the clutter amplitude, or:

(4) seventy-cluttervisibuimEs/A t! is convenient to work with this figure in decibels, kso

at: Y I (s) sov in db='2o1og, s/A

In the signal simulator, as explained above, the vector a is shifted in phase to the position indicated by vector b by an RC phase shifter operating at intermediate frequency. In this phase shifter Xc is kept fixed and the R component is varied by an amount AR, by changing the gm of the doppler modulator. An important factor which must be known and accurately controlled in order to obtain accurate` known amounts of phase shift is the factor A'R/R which gives the fractional decrease in output resistance 'of doppler modulator 24 from the large value of which gives vector b, to the smaller value R-AR which gives vector a. Since the output resistance 1s approximately l/gm for doppler modulator 24 we can say: (6) `f Rei/gm where gm is the lower land Gm the higher transconductance of doppler modulator 24. Dividing, we obtain:

Examination of the expression'for AR/R shows that it represents the fractional increase in transconductance required to bring it from its low value, gm, to its maximum value, Gm, expressed as a fraction of its maximum value. For example, if gm is 70% of the maximum Gm, thenAR/R is 0.30. Using this fact, it is possible to determine the AR/R value, i.e., the change in transconductance required in doppler modulator 24 to shift the signal phase of vector a to 'vector b whatever the positions chosen for vectors a and b.

The geometry of Fig. l1 can be used to compute the AR/R values which correspond to various sub-clutter vislbility values S/A. Assume that vector a is set to the angle 0 with the X, or reactance Axis, and that vector a hasrthecomponent Xc and R-AR along the reactance and resistance axes respectively. As explained above, by decreasing the transconductance of modulator 25, the resistive component is then increased from R-AR to R, producing vector b as the modulator 24 output impedance. We ca ll the angle between vector a and vectorb A0, and therefore vector A, which bisects this angle makes angles 0/ 2 with vector a and vector b. Vector A goes to the midpoint of the line joining the tips of vector b and vector ka (the residue), since vector ka=vector b, and the bisectorof the vertex angle of an isosceles triangle is the perpendicular bisector vofthe base. We can utilize the fact that vector A is perpendicular to the residue vector bysaying: r

n s/Aemn half the angle A0 between vector aV and vector b. To compute still using Fig. 1l, we Write: v (11) tan8=RSfR Substituting for Xe, we obtain:

mn (HM) :1g/565g: talig Solving .for

The angle owe assume knownand fixed byour previous choice of Xc and R-'AR or l/Gm. We canl find the angle. A0 termsl ofY thesubfclutter Visibilityl from theV earlier equation: i

Using (17), we can put Equation 15 in arform allowing computation of ,AR/R in terms of the known quantities (set at 27 degrees), a'ndVS/A, the sub-clutter visibility:

AR 1 133119 `i j R tan (641-2 are tan` S/A) v 4 j From Fig. 11, We canV see that'theratio S/A ywill. be the same, no` matter to what length the two vectors,` along a and b are limited, since the figures so formed will, bythe laws of similar triangles, have"exactlythe'saineproportions, and still yield sub-clutter visibility ratiosequalto S/A. f

Referring again to Fig. 5', there are shown.y calibration circuits which facilitate the calibration off'thefdoppl'er modulator 2'4. TheV square' wave pulse'outp'iit of ipiiop trigger 32 appears also" across cathoderesistor` 1 30 and is used in aY-coniparison" and metering-circuit'.- This square wave pulse has Va repetition rate one-half that of the radar pulse repetition rate and, everifiiieluding the first twenty harmonics requiredA for the squar'efw'ave shape, it does not require a band pass of more than-8 kilo'cycles; Condensers'1v29=and 128r present a relativelyhigh'imped ance to a signal of'this frequency whereas `':oil1"'12f4'Y and movable coil 105 of the phase shifter presentr a'relatively low impedance path to ground.- Theefectiveycathode load for the square'wave pulls'ethereforeV is resistor 130 which in practice had a value of 500 ohms;` Squarefwave pulses taken from the cathode 110 are filtered by; coil 141'- and condenser 142 and'V fed through coupling ca-V pacitor 143 to one endl ofthe primary-v winding 145 of transformer 144. The amplitude of`V these pulses' is aV function of the transconductance of tubelS/arid therefore of its grid voltage. Y

The other end of primary Winding'145 receives a i'efing 145 represents the difference betweemthegvoltages ap.- plied to the respective ends of the primary winding. This difference voltage is coupledfthrough secondary windingy 146 to the control grid 148 of meter amplifier 147. The

amplified difference ,voltage atpthe plate 149 is applied through coupling condenser 151 to rectifier 1,52. The rectifier rectifies thev difference voltage and inthe calibrate l and calibrate 2 position, switch arms 153 and 154 place microammeter 155 in circuit with the rectifier.

Two internal calibration adjustments are provided forV theV doppler modulator circuit. When in` the calibrate l position (switch arm 33 in contact with contact'135) Y resistor 131may be adjusted to set the transconductance of doppler modulator v24 to the desired quiescent operating condition. Clamper diode 136 is disconnected for this adjustment so that the pulses received at the grid 109 1 will have positive as well as negative swings about the grid voltage set'by resistor 131 and thetransconductance :of 5 nautical miles.

i frequency stage 25.

measured will beL the true dynamic value. When the calibration adjustment is made,..doppler switch 34 isset at Steady and arm 138 is set at t-l4.8 db and resisto`r131' i wave output of fiip-liop trigger 32, after having itsl D.C.

component removedv by-condenser 139 and its positive peaksiclamped to the voltage set by resistor 131, causes tube 108 ytooperate for half the time at the quiescent bias voltage and for the other half of the time at a bias yalue determined'by the amplitude of the negative-going signal tapped-off by arm 138. Under'these conditions, the comparison and metering circuit measures the average transt conductance of tube 108, which, since the quiescent transconductance is fixed, is proportional to the transconductance'of the tube for its larger phase angle output. When the calibration adjustment is made, arm 138- is set at -13.1fdb and resistor 114 is varied to obtain a null on microa-mmeter 155.

The detected radar pulses fed to coaxial cable 72 .(Fig. 2) from the plate output of the balanced mixer circuit, in addition tobeing; supplied to the pulse amplifier 30, are also applied across resistor 159 (Fig. 5). When switch 'arms 153, 154 make contact with contacts 160, micro'ammeter' 155jis placed in shunt with resistor 159 aridinthis position may be used to measure the power of the radar input pulses. This is a convenient means for measuring the day-to-day perfor-mance of the radar transmitter; Y

Figf6 is a schematic diagram of the first intermediate- In the embodiment of the invention built, the'intermediate-frequency amplifiers were staggertuned to provide a 2-2.5 megacycle band width with a center frequency at 17.5 megacycles. Y Referring now to Fig. 6, there is shown` a pentode amplifier tube having a control grid 171, screen` grid. 172, suppresser grid 173 and anode 174; The intermediate-frequency pulse output of doppler modulator 24 is supplied tothe suppresser grid 173. The intermediatefrequency output of delay means 27 (Fig. l) is fed to control gridr 171. In a preferred embodiment of the invention, the delay means comprises a quartz delay cell. and, as 'already mentioned, each intermediate frequency pulse applied from the delay cell is delayed the equivalent The gating pulse from one-'shot' multivibrator 31 is applied through Vcrystal diode' 35 to t'hecontroll grid 171 and is used to gate the tube on and Off." Y v In operation, when a positive pulse is supplied to recti* fier 35 it causes the same to conduct and this reduces the 'ilicontrol grid circuit impedance to a low value, eieci tively shorting out the control grid. On the other hand,-

when rectifier 35 is not conducting, that is, during the negative-going portion of the output of 31 (portion 182 of waveform 181 as shown in Fig. 7), the grid impedance is raised to its normal value and tube 170 is in condition to amplify intermediate frequency signals supplied to its control grid. The negative pulse derived from the one-shot multivibrator is applied to rectifier 35 coincident with the arrival of an intermediate frequency pulse from doppler modulator 24. This negative pulse prevents the rectifier from conducting for a length of time sufficient to accumulate the desired number of intermediate-frequency pulses from quartz delay cell 27. At the termination of the operative interval, the gate pulse polarity reverses causing rectifier 35 to conduct and cutoff the intermediate frequency amplifier. This condition is maintained until the next succeeding radar pulse is received.

The gated output of the first intermediate frequency amplifier is capacitively coupled to the control grid circuit of the second intermediate frequency amplifier 26 (Fig. l). The screen grid voltage of the second amplifier is adjustable. This permits control of the gain of the closed loop circuit 25, 26, 27. In a preferred embodiment of the invention, the loop gain is adjusted to unity so that all simulated target pulses have the same amplitude. However, if desired, the loop gain may be varied above or below unity. ln the former case, a series of pulses of ascending amplitude is produced and in the latter case a series of pulses of descending amplitude is produced.

The third amplifier stage 28 is of conventional design and is not discussed here in detail. The output of the third amplifier is applied to monitor detector 36 and attenuator 29. As already mentioned, the detected intermediate frequency pulses may be viewed by connecting a test oscilloscope to jack 38 (Fig. l).

Inra preferred embodiment of the present invention, attenuator 29 compries a so-called piston attenuator which is, in effect, a transformer having a mutual inductance which is determined by changing the physical spacing between the primary coil thereof and the secondary coil thereof. The induced voltage in-the secondary coil is a function of the mutual inductance and therefore by varying the spacing of the coils an output voltage which is variable is obtained. In the embodiment constructed, the output attenuation was variable between and 70 relative db. The output of attenuator 29 is applied to the primary winding 64 of transformer 74 as shown in Fig. 2.

Referring now to Fig. 7, there is shown the time relationship between waveforms at various points in the signal simulator circuit. Pulses 180 represent the received radar pulses. Waveform 181 is the output of oneshot multivibrator 31 and comprises a negative-going portion 182 the duration of which determines the conductive interval of intermediate frequency amplifier 25 and a positive-going portion 183 the duration of which determines the time interval the intermediate frequency amplifier 25 is effectively cut-off. As can be seen,'the leading edge of each negative going square wave portion is coincident with radar pulse '180. The leading edge 184 of waveform 181 triggers flip-flop trigger 32 and causes it to produce an output 185 having a positive going portion 186 and a negative-going portion 187. The positive going portion 186 is clamped by means of diode clamper 136 (Fig. 5) at the level determined by the setting of resistor 131 (Fig. 2). The amplitude of the negative going portion 187 of waveform 185 is adjustable in accordance with the setting of arm 138 (Fig. 5) in the Steady and Pulsed positions.

As already explained, in the Steady position of the doppler switch pulse trains 188, 189, 188a, 189a result. Adjacent ones of the trains are of different amplitudes and different phase. Although each pulse train is shown in the drawing as comprising six pulses, this is merely for' ease of illustration, it being understood that the number' of pulses in each train is variable. Thus, for example, if desired each train may comprise only a single pulse, however, in a prefererd embodiment of the invention each train comprises at least ten pulses. It should also be understood that although in the embodiment described the simulated pulses are spaced the equivalent of 5 nautical miles, it is possible to use any delay value desired.

In the Off position pulse trains 190-1900 result. Adjacent ones of these trains are of the same amplitude and in the same phase.

In the position alternate pulse trains are eliminated leaving pulse trains 191 and 191:1.

The output of the free running multivibrator 33 is'V shown as Waveform 192 including positive-going portion 193 and negative-going portions 194. This waveform and the one following are drawn to a different scale than the preceding waveforms. The free running multivibrator feeds its output to fiip-fiop trigger 32 which causes a pulsed pattern indicated by brackets 195, 196.

Fig. 8 illustrates a plan position (P.P.I.) display (20 mile range) with simulated fixed targets. These targets are indicated by rings 20o-203. In order to obtain such a presentation the signal simulator is set in the doppler Off position. The ground clutter and other fixed targets return signals are lindicated by the cross hatched areas 204.

Fig. 9 shows the same P.P.I. presentation as Fig. 8 under MTI operation and jwith the signal simulator in the doppler Pulsed position. As can be seen, a large portion of the fixed clutter 204 is eliminated. The simulated moving targets are seen as bright spaced dashes 210 which increase in length with increase in range. Although not readily apparent from the diagram, the brightness of the simulated targets decreases with range due to the faster scan rate at increased range. There can also be seen the fixed component 211 between brighter simulated moving targets 210. In areas substantially free of clutter such as 212, vthe simulated targets will be brightest and the best contrast obtained. In areas of clutter the simulated targets will dim or completely disappear. t

In the measurement of subclutter visibility, arm 13S (Fig. 5) is first set at the -10 db position and this causes very bright dashes to occur. The radar P.P.I. cursor is then set to cross a given ring of simulated targets in an area substantially free of clutter at the range desired and arm 138 is moved toward the -40 db position. This causes the very bright dots gradually to decrease in intensity. At the same time the phase shifter 23 (Fig. 5) is set by means of hand crank 107 until maximum contrast is obtained at the desired range. This adjustment should be checked occasionally to compensate for slow changes in the phase of the radar unit. When the db setting is such that the simulated target dots are just barely perceptible against the residue, the db reading is the subclutter visibility of the radar unit under test.

In making the above measurement, it is recommended that the radar antenna speed be so chosen that the best contrast possible is obtained at the desired range. slow a speed, however, causes the dashes to blend together, unless it is possible to increase the repetition rate of the radar unit.

Measurements made at close range do not take into account the rate of sub-clutter visibility deterioration with range. For this reason, it is recommended that a l5 or 20 mile range be used.

Since the simulated target pulses are spaced a fixed distance apart they may be used as a means for checking the accuracy of the range markers. Thus, if the interval between simulated pulses is 5 miles the simulated Too fie

to (if there is a, ixedjd'elay between the occurrence of the receivedy radar pulse and .the first simulated pulse) f the Smile range markers.

As already mentioned, the frequency of the stable local oscillator may be controlled. This Aenables one readily to check the radar frequency. i Inthe embodiment actually built, the stable local oscillator frequency control included a counter dial which indicated the exact setting of the frequency control adjustment. setting. of the stable. local oscillator frequency is a function of the output frequency of the radar unit under test, it is possible to draw up a frequency correction chart relating the stable local oscillator'frequency to the radar output frequency and itis thereby possible to detect changes in the radar output frequency and vely easily to correct the same. Y.

Since the Y i6 intended to be comprehended within the meaning and rangev of equivalence of the following claims. I

What is claimed 4asl'rreyvand desired to; be secured by Letters. Patent f is :v

Itwas mentioned above in connection with themeasurement' of Vrange marker accuracy that the train, of simulated target pulsesirnay be delayed in theirfstartl by a predetermined amount relative to the start of the radar pulse. lnthe unit built,. the delay inserted wasV 1/2 microsecond. Accordingly, the first simulated targetV pulse should appear about 80 yards after theY radar transmitter pulse and succeeding pulses should appear 80 yards after their corresponding range markers. This beats between the radar transmitter pulse' and the firstsimulated target pulse can be seen on the radar Aoscilloscope atopl the radar transmittepulse.

In the 100% doppler position, a set of coherent targets is transmitted back to the radar receiver every other pulseV period. This form of signal may be considered as pure moving target with no fixed component. ,By scanning ythe P.P.I. oscilloscope with this signal andreducingA the output level of the simulator until the MTI signal trace is just discernible in clutter, the minimum detectable moving target signal V(as referred to. clutter level or to receiver noise level) may be determined.

Although a number of specific exampleshave been given of the use of :the present invention, it is quite apparent that many other uses of the invention are possible, the abolve examples being given only by way of illustration.

It will be understood that each of the elements described above, or two or more together, may also find a useful application in other types of signal simulators differing from the types described above.

While the invention has been illustrated and described as embodied in a signal simulator especially adapted for use in connection ywith MTI type radar units, it is not intended to be limited to the' details shown, since various modifications and structural changes may be made without departing in anyway from the spirit of invention.

Without furtherV analysis, the foregoing will so fully reveal the gist of the present invention that others can by applying current knowledge readily adapt it for various applications without omittingV features that, from the standpoint of prior art, fairly constitute essential characteristics of the generic or specific aspects of this invention' and, therefore, such adaptations should andV are If the radar transmitter pulse length isV the present 1. A radar signalL sinjulatorfV comprising, in combina-V tion, receiver means adapted tobecoupled to a radar unit for receiving carrier frequency radar pulses; hetero'- y dyningVV means coupled to said receiver means for converting received carrier frequency radar pulses to intermediate frequency pulses; generating means coupled to said heterodyning means for producing a` group of retardedV intermediate frequency pulses in synchronism With` each of said intermediate frequency pulses respectively, respective pulses in each of said groups being spaced in time from one anotherby the same predetermined. interval of time; phase shifting means coupled to said generating means for shifting the phase of the intermediate fre# quency of adjacent groups of retarded intermediate frequency pulses by a predetermined amount and'for main-V taining the phase of the intermediate frequency ofr'alternate of said groups of retarded intermediate frequency pulses the same, said phase shifting means comprising a cathode follower having a capacitance cathode load imvalues in timed relationwith the reception of, said carrier frequency pulses; and means coupled to said phase shifting means for converting said groups of retarded in# termediate frequency pulses to said carrier`frequency.

2. A radar signal simulator comprising, in combination, receiver means adapted to be coupled to a radar unit for receiving carrier frequency radar pulses; heterodyning means coupled to said receiver means for con- 0 verting received carrier frequency radar pulses to intermediate frequency pulses; generating means coupledA tol said heterodyning means for producing a group oferetarded intermediate frequency pulses in synchronism with eachy of saidintermediate frequencyV pulses respectively, Y

ing means forrshifting theV phase of the intermediate fre-l quency of adjacent groups of retarded intermediate frequency pulses by a predetermined amount and for maintaining the phase of the intermediate frequency ofalterf nate of said groups ofl retarded intermediate frequency pulses the same, said phase shifting means4 comprising a' means coupled to said'square wave generating means synchronizing' the same to the reception of said carrier lfrequency pulses; and means coupled to said phase shifting means for converting said groups of retarded intermediate frequency pulsesto said carrier frequency.

3. A radar signal simulator comprising; in combination, receiver means adapted to be coupled to a radar unit for receiving carrier frequency radar pulses; heterodyning means coupled to said receiver means for converting` received V'carrier frequency radar pulses to intermediate'V frequency pulses; adjustablev phase shifting means coupledto Said heterodyning means for shifting the phase 0f said :intermediate frequency pulses Vby an adjustable amount; generating means coupled to said adjustable phaseshifting means for producing a group of retardedV intermediate frequency. pulses insynchronism with each of said intermediate. frequency pulses respectively, re-V spectiveV pulses in each of saidrgroups being spaced in timey from one anothery by the same predetermined interval of. time; phase shifting means coupled to said generatingg means for shifting the phase of the intermediate fre- References Cited in the iile of this patent quency of adjacent groups of retarded intermediate fre- UITED STATES PATENTS quency pulses by a predetermined amount and for mainfrequency, 15 2,841,785 Cunningham et al. July 1, 1958 

